1. Field of the Invention
This invention relates to controllers for loads of varying inductance, such as switched reluctance machines. The invention also relates to a hysteresis circuit.
2. Description of Related Art
A switched reluctance motor is an example of an electrical load in which the inductance is not constant with time or machine operation, i.e. rotor angle. FIG. 1(a) illustrates a typical three-phase switched reluctance (SR) machine and FIGS. 1(b) and (c) illustrate well-known examples of electronic switching circuits which may be used to control an SR machine. The SR machine essentially consists of a stator s defining stator poles 1, 1', 2, 2', 3, 3' on which are wound phase windings w, of which only one is shown in association with a set of poles 2, 2'. The machine also has a rotor r with salient poles 4, 4' and 5, 5'. The electronic circuits are arranged to supply unidirectional currents to the phase windings w.
In the control circuits of FIGS. 1(b) and (c), each phase winding of the machine in FIG. 1(a) is associated with a circuit leg comprising at least one electronic switch t in series with each winding across a dc supply Vs. A general treatment of the principles of SR machines is given in the paper `The Characteristics, Design and Applications of Switched Reluctance Motors and Drives` by Stephenson et al. presented at PCIM'93 Conference and Exhibition at Nurnberg, Germany, Jun. 21st-24th 1993.
Proposed means for controlling an SR machine to run at different speeds include operation in three characteristic regions which can be designated as `low-speed`, `medium-speed` and `high-speed` regions. For background and explanation on this see `Variable-Speed Switched Reluctance Motors` by Lawrenson et al. IEE Proceedings Part B, Vol.127, No.4, July 1980. In the low-speed region the current is controlled by the well-known method of `chopping`. The present invention relates to improvements in the means of implementing chop control of the phase current.
FIGS. 2(a), (b) and (c) represent one phase leg of the circuit shown in FIG. 1(b) redrawn for clarity with the phase leg in its three possible states. Here referred to as `ON`, `FW` (`freewheeling`) and `OFF`. The power switches shown (referred hereinafter as `top switch` and `bottom switch`) may be, for example, bipolar, insulated gate or MOS field effect transistors, gate turn-off thyristors or other power switch means, as is well understood in the art. The power switches are assumed to be operated by suitable driving circuits, as is usual in the art--for clarity these are not shown in the figures. The controller diagrams (FIGS. 4 and 6) therefore show command signals for top and bottom power switches, which are assumed to be transmitted to the switches via the aforementioned driving circuits. The circuit is in the ON state when both power switches are closed and the full dc bus voltage Vs is applied to the phase winding w, increasing the flux .PHI. at the maximum possible rate (see FIG. 2(a)).
When flux has been established in the winding in this way and either one of the switches is open, the current is said to freewheel (see FIG. 2(b)), i.e. the circuit is in the FW state with only one switch closed, the current flowing through this switch and one diode. The effective winding voltage is determined by the small voltage across the switch, the diode and the winding resistance. The flux .PHI. thus falls relatively slowly, as shown in FIG. 2(b).
The circuit is in the OFF condition when both switches are open and the phase current is carried by the diodes. The winding then has the full dc bus voltage applied in reverse so that the flux .PHI. will fall until the current is zero and the diodes become non-conducting (see FIG. 2(c)).
The simplest method of current chopping is to alternate between ON and OFF states to maintain the mean current level near a desired value. This is shown in FIG. 3(a). Chopping between the ON and OFF states is useful at low power levels where the switches (e.g. semiconductor switches such as metal oxide silicon dioxide field effect transistors or insulated gate bipolar transistors) can switch at ultrasonic frequencies. This is advantageous in terms of limiting acoustic noise.
At higher power levels however, the losses (both in the semi-conductor switches and other components) associated with ultrasonic switching become large and it is usually necessary to reduce the switching frequency. If the ON/OFF strategy were used the current (and the flux) excursions at these reduced frequencies might be large, resulting in a rise in objectionable acoustic noise and, possibly, control problems as well. For these reasons the FW state is often incorporated into the switching pattern, enabling retention of relatively small current excursions even though the switching frequency is reduced. This is shown in FIG. 3(b) for a `motoring` mode and in FIG. 3(c) for a `generating` mode.
The behaviour of the winding current is determined not only by the applied voltage but also by the phase inductance which is a function of the rotation of the machine. Inductance is defined as flux linkage per unit current so that L.varies..PHI./I, hence I.varies..PHI./L. If the effective winding voltage during freewheeling is small, the flux can be considered constant over a short period and the current will follow the reciprocal of the inductance profile, i.e. the freewheeling current will reduce when the inductance rises and increase with falling inductance. At low rotational speeds, however, the rate of change of inductance with time (dL/dt) is small and the reduction in flux with time during freewheeling becomes significant over a machine phase period. At very low speeds, the freewheeling current will fall even when the inductance is decreasing, because a small winding voltage causes the flux to fall faster than the inductance. The behaviour of the freewheeling current under these differing conditions has important consequences for the current control system as the controller has to be able to function correctly in all these different conditions.
In the prior art, the control considerations described above have been implemented in a variety of ways, each with its own advantages and disadvantages.
In the prior art, simple current controllers using freewheeling do not cater to the case where the phase current rises during freewheeling (e.g. MacMinn, U.S. Pat. No. 4,933,621). This is acceptable where the switched reluctance machine is never required to operate as a regenerative brake. It is inadequate, however, when attempting to control the phase current during regenerative braking, because under these conditions the machine's phases will be energized while their inductance is falling, which (as explained above and shown in FIG. 3(c)) results in a rising freewheel current at all but the lowest rotational speeds. With the MacMinn controller, the freewheel current under regenerative conditions would rise in an uncontrolled manner, resulting in unacceptable performance of the machine and/or possible damage to the power electronics and/or motor windings.
A known extension of the simple scheme is shown in FIG. 4, where additional control logic is used to modify the power switch commands when the switched reluctance machine is known to be operating in the braking mode. A conventional hysteresis controller is shown, using one comparator for each phase, the current feedback being compared directly with the reference (demand) value. A hysteresis band separates the switching points. The width of this band is usually varied to provide a suitable compromise between current excursions, switching frequency and acoustic noise.
In a motoring mode (i.e. in which the net power flow is from the source to the load), the controller chops simply by alternating between ON and FW, with both switches switched off at the end of the phase period. FIG. 5(a) shows a typical current waveform. The change from motoring to a generating mode requires the MOT/GEN logic signal shown in FIG. 4. In the generating mode (i.e. in which the net power flow is from the load to the source), the controller chops by alternating between FW and OFF. FIG. 5 (b) shows a typical current waveform.
While this system has the advantages of using only one comparator and only one hysteresis band, it has a number of disadvantages. These are particularly apparent at low-speeds whilst generating. Because the generating freewheel current rises only if the speed is high enough, at low speeds (particularly at high currents when the effective freewheel winding voltage is greatest) the current decays to zero as illustrated in FIG. 5(c). The output of the generator is then reduced. A further drawback is the dependence on a logic signal for switching between motoring and generating modes. This signal may be difficult to generate reliably especially during transient conditions. This may lead to loss of control of the current resulting in nuisance tripping or even switch failure.
A second proposed system seeks to overcome these difficulties by using two comparators. The two comparators have the same reference, but the hysteresis band of one spans that the other, as shown in FIG. 6(a). Essentially, the `outer` comparator is used to control the operation of the `inner` comparator so that, for most of the time, chopping occurs between the switching points of the inner comparator only. The power switches are switched off at the end of the phase period as before. In the motoring mode, this system behaves like the single comparator case and chopping is controlled by the inner hysteresis band only, as shown in FIG. 6(b).
In the generating mode, the controller starts in the same mode with both switches closed until the current reaches the upper level of the inner comparator, whereupon one switch is opened and the phase freewheels. The freewheel current rises further (because the system is generating) until the upper level of the outer comparator is reached whereupon the second switch is also opened. The inner comparator now operates in the generating mode, selecting either FW or OFF. Chopping continues on the inner band unless the current fails to rise when the circuit is in the FW state and decays to the lower level of the inner comparator. In this case, if the current falls to the lower level of the outer comparator and both switches are closed, the resulting ON state raises the current into the control band of the inner comparator. This is illustrated in FIG. 6(c).
This system has some advantages. For example, it keeps control of the current at all times, irrespective of whether the drive is motoring or generating and has no problems of discontinuities at low speeds.
However, it has disadvantages which may render it of little value in some applications. It suffers from large transient excursions of current which may be objectionable in some applications. Also, the outer band must generally be wide so as to keep it reliably distinct from the inner one over the full working range of current despite noise, drift or other possible sources of signal corruption.